Comparator low power response

ABSTRACT

In described examples, an amplifier can be arranged to generate a first stage output signal in response to an input signal. The input signal can be coupled to control a first current coupled from a first current source through a common node to generate the first stage output signal. A replica circuit can be arranged to generate a replica load signal in response to the input signal and in response to current received from the common node. A current switch can be arranged to selectively couple a second current from a second current source to the common node in response to the replica load signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.16/378,526, filed Apr. 8, 2019, which is incorporated by referenceherein in its entirety.

BACKGROUND

Electronic circuits are designed to include increasingly smaller designfeatures. The smaller design features of the electronic circuits can beused to attain smaller form factors, increased functionality, andreduced power consumption of the electronic circuits. Such electroniccircuits can include amplifiers (including comparators) for controllingvarious systems. Some comparators are arranged as parts of controlcircuits. The stability and accuracy of such control circuits oftendepend on the latency of the included comparators. However, lowering thepower consumption of the circuitry that includes the comparators canincrease the delay and/or decrease the accuracy of an output signalgenerated by an included comparator.

SUMMARY

In described examples, an amplifier can be arranged to generate a firststage output signal in response to an input signal. The input signal canbe coupled to control a first current coupled from a first currentsource through a common node to generate the first stage output signal.A replica circuit can be arranged to generate a replica load signal inresponse to the input signal and in response to current received fromthe common node. A current switch can be arranged to selectively couplea second current from a second current source to the common node inresponse to the replica load signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of an example comparator for low-powerresponse to input signal fluctuations.

FIG. 2 is a waveform diagram of an example simulation of a disabledlow-power response of the example comparator to a large input signaltransition.

FIG. 3 is a waveform diagram of an example simulation of a low-powerresponse of the example comparator to a large input signal transition.

FIG. 4 is a waveform diagram of an example simulation of a disabledlow-power response of the example comparator to a small input signaltransition.

FIG. 5 is a waveform diagram of an example simulation of a low-powerresponse of the example comparator to a small input signal transition.

FIG. 6 is a waveform diagram of another example simulation of a disabledlow-power response of the example comparator to a large input signaltransition.

FIG. 7 waveform diagram of another example simulation of an enabledlow-power response of the example comparator to a large input signaltransition.

FIG. 8 is a schematic diagram of another example Aux_bias generator forlow-power response of the example comparator.

FIG. 9 is a flow diagram of an example method for a response of anexample low power comparator to input signal fluctuations.

DETAILED DESCRIPTION

Electronic circuits can include control circuits. For example, a controlcircuit can generate a control signal in response to a feedback signal.The feedback signal can be generated by measuring (e.g., comparing) asignal generated in response to a quantity developed at least in part inresponse to the control signal.

In some electronic circuits, a feedback signal can be generated byamplifying a signal (e.g., amplifying a voltage difference betweenrespective conductors of a differential signal) generated in response toa quantity that is generated in response to the control signal. Thestability and accuracy of such control circuits depend on the latency(e.g., delay) of the circuitry for generating the feedback signal.Generally, lowering the power consumption of the circuitry forgenerating the feedback signal can increase delay times and/or decreasethe accuracy of the feedback signal.

In contrast, increasing the power consumption of the circuitry forgenerating the feedback signal can decrease delay times and/or increasethe accuracy of the feedback signal. However, increasing the powerconsumption of the circuitry for generating the feedback signal canresult in decreased operating characteristics, for example increasedpower consumption and increased heat dissipation (e.g., which couldrequire remedial cooling), larger components (e.g., for greater powerratings), decreased battery life (e.g., which could otherwise requiregreater energy storage), and/or increase the need for line power foractive cooling.

Some electronic systems can include amplifiers (e.g., which can includeat least one transistor) that can be arranged for actively controlling asource current in response to input signal voltages. Some examples ofthe amplifiers can be arranged as comparators. For a given sourcecurrent (e.g., for powering a comparator), comparators generally respondquickly to smaller voltage changes of the first and second inputsignals. Also, due to the time during which a common node settles to anappropriate (e.g., ideal) value, comparators generally respond moreslowly to larger voltage changes that can be present in the first andsecond input signals. Increasing the source current output of a currentsource for powering the comparator can increase the speed and/oraccuracy of the comparator. However, increasing the source current ofthe comparator also raises the power consumption of the comparator.Raising the power consumption of the comparator can render a comparatordesign as being unsuited for at least some lower power systems.

An example comparator described herein can be arranged for comparing afirst input signal voltage against a second input signal voltage, andfor generating an output signal (e.g., a single-ended or a complementaryoutput signal) in response to the comparison. The example comparator canselectively increase power during some input conditions that couldotherwise degrade performance (e.g., increase latency and/or decreaseoutput accuracy).

FIG. 1 is a schematic diagram of an example comparator 100 for low-powerresponse to input signal fluctuations. The comparator 100 includes afirst stage 110, replica input transistor pair 120, an Aux_biasgenerator 130, current switch 140, a second stage 150, and a third stage160. In at least one implementation, the comparator 100 is arranged tocompare a pair of differential input signals and to selectively addcurrent to a common node of a differential amplifier when acurrent-starved condition of the common node is detected.

The first stage 110 is an amplifier having an input transistor pair 112and a current mirror 115. The input transistor pair can comprise PMOS(P-type metal-oxide-semiconductor) transistors such as transistors Q1and Q2, where the transistors Q1 and Q2 include a common node (e.g.,common source node 114) coupled to at least a first current source I1.In other examples, transistors Q1 and Q2 are different types oftransistors. The current mirror includes NMOS (N-typemetal-oxide-semiconductor) transistors Q3 and Q4, and resistors R1 andR2 (e.g., for generating the Replica load CG signal for biasing thecommon gates of Q3 and Q4). The drains of Q3 and Q4 are respectivelycoupled to the drains of Q1 and Q2. The resistors R1 and R2 (e.g.,arranged as a voltage divider) are coupled in series between therespective drains of Q3 and Q4 (e.g., where the drains are coupled asfirst and second inputs of a current mirror that includes Q3 and Q4). Inother examples, transistors Q3 and Q4 are different types oftransistors. A central node 113 (e.g., divided voltage node) between theresistors R1 and R2 is coupled to bias respective control terminals(e.g., to commonly bias the gates) of Q3 and Q4.

The first stage 110 is a differential amplifier coupled todifferentially receive input signals, such as a positive input signal(INP) and a negative input signal (INM). The signal INP and the signalINM respectively control currents flowing through Q1 and Q2. Asdescribed hereinbelow, the voltage of the common source node 114 isdeveloped in response to the current sourced by a first current source(e.g., current source I1), and in response the currents selectivelycontrolled by transistors Q1 and Q2. In some conditions (e.g., resultingfrom voltage changes of the input signals), a current-starved conditionof the common source node 114 can develop, so that the voltage rise ofthe common node can be delayed and/or result in erroneous output of thecomparator 100.

The replica input transistor pair 120 is a replica circuit of the inputtransistor pair 112. For example, the replica input transistor pair 120includes PMOS transistors Q5 and Q6, which can be the same size orotherwise scaled to determine (e.g., detect, imitate, emulate, and/orsimulate) a performance (e.g., at least one operational characteristicof Q1 and/or Q2) of the input transistor pair 112. The source nodes ofQ5 and Q6 are coupled to the common source node 114, and the gates of Q4and Q6 are respectively coupled to signals INP and INM. In such anarrangement, the replica input transistor pair 120 can respond to thesame (or similar) contemporaneous input conditions to which the inputtransistor pair 112 is subjected. For example, the replica inputtransistor pair can detect a voltage drop of the common node, where thedetected voltage drop is generated in response to a rise of an inputsignal (e.g., signal INP or INM).

The output of the replica input transistor pair 120 (e.g., the commonlycoupled drains of Q5 and Q6) is a replica signal (e.g., Replica_load)for indicating (e.g., emulating) a contemporaneous response of the inputtransistor pair 112. As described hereinbelow, the replica signal can becoupled along a feedback path to generate a feedback signal (e.g.,Aux_bias) for controlling the herein-described selective addition ofcurrent to the common source node 114 via the current switch 140. Duringa current-starved condition, for example, the output current (e.g., tailcurrent) of the replica input transistor pair 120 is decreased, whichindicates the current-starved condition. In response to the decrease ofthe tail current of the replica input transistor pair 120, the Aux_biasfeedback signal is asserted, so that current flowing through the drainof Q7 contributes supplemental charge to the common source node 114.

The Aux_bias generator 130 is coupled to receive the output of thereplica input transistor pair 120. The Aux_bias generator 130 includes acurrent source I2, an NMOS transistor Q8, and an NMOS transistor Q9. Thetransistor Q8 is biased by a normalized cascode (ncas) control signaland Q9 is biased by a normalized bias (nbias) control signal. Therespective control signal voltages are selected, so that the Aux_biasgenerator 130 asserts the Aux_bias signal in response to a decrease inthe current of the Replica_load signal (and de-asserts the Aux_biassignal when the Replica_load signal indicates the current-starvedcondition of the common source node 114 has been reduced).

When the signal Replica_load indicates a current-starved condition ofthe common source node 114, less current is added to the current flowingthrough the drain of Q9 (where the current otherwise flowing through Q9is coupled from I2 via Q8). In response to less current being added tothe current flowing through Q9, the voltage (e.g., of node Aux_bias) ofthe source of Q8 falls, so that PMOS transistor Q7 is turned on.

The current switch 140 includes Q7 and the resistor R3. Resistor R3 is acurrent source for coupling a limited current to be selectively coupledthrough Q7 into the common source node 114. The node Aux_bias is coupledto the control terminal (e.g., gate) of Q7. The transistor Q7 isarranged to selectively apply current to the common source node 114 inresponse to the Aux_bias voltage. In at least one example, the currentsource I1 is first current source, the current source R3 (e.g., which iscoupled to VDD power rail) is a second current source, and the first andsecond current sources are coupled in parallel between a power rail andthe common node.

The second stage 150 is a second stage amplifier that includes PMOStransistors Q10 and Q11 and NMOS transistors Q12 and Q13. The secondstage amplifier is coupled to convert a differential input received fromthe first stage amplifier to single-ended output of the second stageamplifier. The transistor Q10 is a master transistor and the transistorQ11 is a slave transistor. The master and slave transistors are arrangedas a current mirror for generating a second stage output signal inresponse to a first stage output signal (e.g., first stage outputsignals that are differential). For example, the current mirror (Q10 andQ11) transistors are biased in response to the first stage output-minussignal (1st_stage_out_minus signal), while the second stage outputsignal (2nd_stage_output signal) is developed in response to the1st_stage_out_plus signal and in response to the current supplied byQ11. The second stage 150 can be arranged as a differential input tosingle-ended output converter. In another example (not shown), thesecond stage 150 can be arranged having a differential output, so thatthe comparator 100 can be arranged having a differential output.

The third stage 160 is an amplifier (e.g., buffer and/or output stage)arranged to quantize the second stage output (e.g., an analog signal,2nd_stage_output), and to output a signal (e.g., digital signal) forindicating the result of the comparison of the pair of differentialinput signals. For example, the third stage can include an odd number ofinverters for buffering and inverting the second stage output togenerate the output signal as a comparator output (Comp_Out).

In examples described herein, a low-power response comparator includesan input transistor pair (e.g., input transistor pair 112) arranged forreceiving first and second input signals (e.g., voltages that can varyover time). The input transistor pair can include a first transistor(e.g., Q1) and a second transistor (e.g., Q2). A first current terminal(e.g., source or drain) of the first transistor is coupled to a firstcurrent terminal of the second transistor and to a common node (e.g.,common source node 114). A control terminal (e.g., gate) of the firsttransistor is coupled to the first input signal, and a control terminalof the second transistor is coupled to the second input signal.

A first current source (e.g., always-on current source I1) includes acurrent output coupled to the common node, so that (for example) thefirst current source provides a first current coupled through the commonnode to the respective current terminals of the first and secondtransistors. (The term “source” need not refer to a source terminal of aPMOS or a NMOS transistor and can refer to a source of a positive ornegative current depending on context.)

A transistor of the input transistor pair can individually control arespective current in response to an input signal coupled to a controlterminal of the transistor of the input transistor pair. In an example,each transistor of the input transistor pair can be arranged toindependently control (e.g., by varying a current carried between sourceand drain terminals in response to a control signal) a portion of thesource current that is sourced from the common node of the inputtransistor pair.

In an example, the respective sources of the input transistor pair arecoupled to the common node, and the respective drains of the inputtransistor pair are coupled to respective drain nodes. Accordingly, avoltage change of a respective first and second input voltages cancontrol the respective magnitude of the first and second currents, wherethe first and second currents respectively flow from a source to a drain(or, for example, from a drain to a source) of a respective transistorof the input transistor pair.

An example low-power response comparator (e.g., comparator 100)described herein can include a replica circuit (e.g., replica inputtransistor pair 120 circuit) for indicating (e.g., replicating oremulating) a response of the input transistor pair to changes in thefirst and second input signals. The replica circuit can include areplica input transistor pair that is coupled to receive a current fromthe common node that sources a current to the input transistor pair. Thereplica input transistor pair need not be instantiated (e.g., physicallymanufactured) using the same exact design features of each transistor ofthe input transistor pair; for example, the transistors of the replicainput pair can be scaled, so that the replica circuit can emulate ascaled response of the input transistor pair.

An example replica input transistor pair can include a third and afourth transistor. A first current terminal of the third transistor iscoupled to a first current terminal of the fourth transistor and to thecommon node. A second current terminal of the third transistor iscoupled to a second current terminal of the fourth transistor and to afeedback signal node. A control terminal of the third transistor iscoupled to the first input signal, and a control terminal of the fourthtransistor is coupled to the second input signal.

As described herein, the replica input transistor pair can emulate theinput transistor pair. For example, the replica input transistor paircan generate a replica signal (e.g., at a Replica_load node of FIG. 1)for indicating (e.g., emulating) a contemporaneous response of the inputtransistor pair. The replica signal can be arranged to generate afeedback signal (e.g., Aux_bias signal of FIG. 1) for indicating acurrent-starved condition of the common node that is generated by theinput transistor pair (e.g., generated at the common source node 114).

The replica input transistor pair can detect (e.g., by emulation of theinput transistor pair) the current-starved response of the inputtransistor pair because, for example, the replica input transistor pairis coupled to like inputs (or buffered inputs derived from the likeinputs) that are coupled to control the first transistor pair (e.g.,input transistor pair).

The current-starved response is a response of the input transistor pair(e.g., input transistor pair 112) to the current sourced by the firstcurrent source and to the first and second input signals. For example, acurrent-starved condition can exist (e.g. result or develop) in a nodebetween two transistors where the first transistor supplies a current tothe node, and the capacitance of the node (and current conducted awayfrom the node by the second transistor) impedes a voltage change of thenode.

To determine the current-starved response, the second current terminalsof the third and fourth transistors are coupled to a feedback signalnode to generate a combined current. The combined current is anindication (e.g., the Replica_load signal) of the determined performanceand is coupled to the feedback signal node for generating a feedbacksignal (e.g., the Aux_bias signal) for controlling a current switch(e.g., current switch 140).

The current switch includes a first current terminal coupled to a powerrail, a second current terminal coupled to the common source node, and acontrol terminal coupled to the feedback signal node. The current switchis arranged to selectively couple current into the common node inresponse to the indication of the determined performance. A biasgenerator (e.g., Aux_bias generator 130) can generate a feedback signal(e.g., bias signal) in response to the indication of the determinedperformance. The feedback signal controls (e.g., activates and/orregulates) the current switch (e.g., a boost current source such ascurrent switch 140). For example, the bias generator is arranged toassert the bias signal to activate the current switch in response to adecrease in current of the indication received from the feedback signalnode.

Accordingly, a feedback loop exists, so that (for example) the currentswitch regulates (e.g., selectively provides in response to the feedbacksignal node) the coupling of a second current (e.g., augmentationcurrent) to the common node. The feedback signals can include signalsdeveloped in the associated feedback path, such as the control signalfor selectively controlling the current switch.

The current switch can be coupled to generate (e.g., amplify and/orinject) a controlled boost current for dynamically augmenting the sourcecurrent for powering the input transistor pair. The source current forpowering the input transistor pair can be selectively applied bycontrolling the addition of a boost current (e.g., as regulated by thecurrent switch) to the common source node of the input transistor pair.Selectively augmenting the source current for powering the inputtransistor pair can save power that would otherwise be expended byproviding, for example, a fixed magnitude source current (e.g., of thefirst current source) to avoid a current-starved response of the commonsource node of the input transistor pair.

The augmented source current is coupled to the first and second currentterminals of the input transistor pair. In an example configuration, asecond current terminal (e.g., drain or source) of the first transistorcan be coupled to a first input of a second stage of the comparator anda second current terminal of the second transistor can be coupled to asecond input of a second stage of the comparator. In the exampleconfiguration, the output signal (e.g., analog output signal) of thesecond stage can be quantized (e.g., converted to a digital value) andcoupled as an output signal (e.g., as digital output) of the comparator.As described herein, the selective augmentation of the commonnode-sourced current can help reduce latencies and output errors whilemaintaining low power consumption of the low-power response comparatordescribed herein.

The selective augmentation of the common node-sourced current forpowering the input transistor pair can reduce static currents otherwiseconsumed by the input transistor pair and reduce latency (e.g., asdescribed herein with respect to FIG. 1, FIG. 3, FIG. 5, and FIG. 7).Moreover, the selective augmentation of the source current can increaseamplifier accuracy and eliminate some kinds of spurious comparatorerrors (e.g., as described herein with respect to FIG. 6).

A current-starved response of the common source node of the comparator100 can be shown by disabling the turning on of the current switch 140in simulations. For example, when the current switch 140 is disabled insimulations, latencies resulting from a current-starved common sourcenode are shown (e.g., as described herein with respect to FIG. 2, FIG.4, and FIG. 6), and an output error resulting from a current-starvedcommon source node is shown (e.g., as described herein with respect toFIG. 6). The simulations were performed using a simulation program suchas SPICE (simulation program with integrated circuit emphasis) tomathematically generate responses of a modeled circuit to input signalfluctuations.

FIG. 2 is a waveform diagram of an example simulation of a disabledlow-power response of the example comparator to a large input signaltransition. The example simulation 200 includes waveforms for showing anexample operation of portions of the comparator 100, describedhereinabove with reference to FIG. 1. The example waveforms include thewaveform INM 210 (e.g., the “input minus” signal coupled to the gate ofQ2 of FIG. 1), the waveform INP 220 (e.g., the “input plus” signal ofQ1), the waveform Comp_Out 230 (e.g., the “comparator output” signal ofthe third stage 160), the waveform Source 240 (e.g., of the nodeCommon_Source_Node coupled to the respective sources of transistors Q1,Q2, Q5, and Q6 and coupled to the drain of Q7), and the waveformAux_bias 250 (e.g., coupled to the gate of Q7). The low-power enhancedresponse of the example comparator to a large input signal transitioncan be disabled in response to a simulation parameter (e.g., a feedbacksignal to generate the low-power response can be disabled by couplingthe Aux_bias node to 3.9 volts via an ideal switch in the simulation200).

In the simulation 200 for illustrating a current-starved response of thecomparator 100 to a large input signal transition, the waveform INM 210is initially asserted at a ground potential (e.g., 0 volts) and thewaveform INP 220 is initially asserted at around 2.90 volts. Because themagnitude of the waveform INP 220 is greater than the magnitude of thewaveform INM 210 (e.g., under steady-state conditions), the waveformComp_Out 230 is initially a logic one (e.g., a logic high level, whichis represented here as a voltage greater than 1.42 volts, for example).

The waveform Source 240 indicates the voltage of the common source node114 is initially driven to around 1.2 volts in response to the currentsourced by a first current source (e.g., current source I1), and inresponse the currents selectively controlled by transistors Q1 and Q2.For example, the waveform Source 240 is driven to around 1.2 volts inresponse to the waveform INM 210 being at ground potential (e.g., whichstrongly biases Q2 to conduct), in response to the waveform INP 220being at around 2.90 volts (e.g., which moderately biases Q1 toconduct), and in response to the commonly controlled current mirror ofQ3 and Q4 (e.g., which are commonly biased via the resistor network ofR1 and R2).

The waveform Aux_bias 250 is initially driven to a value of around 3.90volts in response to the current source I2. The low-power enhancedresponse of the example comparator to a large input signal transitioncan be disabled in response to a simulation parameter (e.g., a feedbacksignal to generate the low-power response can be disabled by couplingthe Aux_bias node to 3.9 volts via an ideal switch in the simulation200).

During operation of the comparator 100, the feedback signal of thereplica input transistor pair 120 circuit can be a voltage developed inresponse to the “tail” current of the replica input transistor pair 120circuit. In the simulation 200, the feedback signal is decoupled (e.g.,as a function of a simulation 200 input parameter) from the waveformAux_bias 250 (when the transistor Q8 is in the off state). The feedbacksignal is disabled (e.g., by turning off Q8 in response to a simulation200 input parameter), so that, for example, a response of the comparator100 with a disabled low power enhancement can be seen. Without thedescribed low power selected current-boost enhancement being enabled inthe example, the simulation 200 of the comparator 100 shows a longlatency (e.g., around 230 nanoseconds) of the voltage (e.g., waveformSource 240) of the common source node 114 rising to a steady-statelevel.

The waveform Aux_bias 250 is coupled to the control terminal (e.g.,gate) of the transistor Q7. The transistor Q7 is arranged as aprogrammable (e.g., programmable in response to a gate voltage) currentsource for selectively applying current to the common source node 114.In the simulation 200, the transistor Q7 is biased (in response to asimulation 200 input) against selectively applying current to the commonsource node 114 (e.g., applying current in response to the feedbacksignal generated by the replica input transistor pair 120 circuit).Because the waveform Aux_bias 250 is around 3.90 volts, the PMOStransistor Q7 is in the off state, so that no boost current is injectedby the current source Q7 into the common source node 114 (of Q1 and Q2,for example).

At 10 microseconds into the simulation 200, the waveform INM 210 isdriven (e.g., as a simulation 200 input parameter) to undergo a largevoltage transition 212 from a ground potential to a voltage around 2.92volts (e.g., which is close to—but greater than—the contemporaneousvoltage of the waveform INP 220). The transition 212 of waveform INM 210causes transients 222 and 252 (for example, via parasitic couplingand/or “ground bounce”).

In the response to the transition 212, the gate voltage of transistor Q1is raised to around 2.92 volts. Accordingly, the gate voltage of Q1(after transition 212) is higher than the contemporaneous gate voltageof Q2. The current source I1 is designed to source current at a lowmaximum (e.g., by design, to save power). The low level of the maximumcurrent can result in a current-starved response, which contributes tothe latency (e.g., delay) of the voltage rise (during transition 242) ofthe common source node 114. The waveform Source 240 during transition242 is raised (e.g., slowly) in response to the limited current that issourced by the current source I1, the common source node 114capacitance, and the current drained by the current mirror that includestransistors Q3 and Q4.

Parasitic conditions of the structures forming the common source node114 (e.g., of Q1 and Q2) impede a rise (e.g., instantaneous rise) in thevoltage of the common source node 114. Accordingly, the slew rate of thetransition 242 is limited, and the rise of the waveform Source 240 toaround 3.6 volts is achieved with a latency of around 200 nanoseconds.(In the simulation described hereinbelow with respect to FIG. 3, alow-power enhanced response can decrease the latency of the voltage riseof the common source node 114 by 140 nanoseconds for similarcharacteristics of the waveform INM 210 and the waveform INP 220).

In the simulation 200, the waveform Comp_Out 230 toggles (e.g., toggleslow) in response to the transition 212 of the waveform INM 210 to avoltage greater than the contemporaneous voltage of the waveform INP220. In response to the transition 212, the waveform Comp_Out 230toggles from a logic one to a logic zero (where a logic zero isrepresented as a ground voltage and a logic one is represented as alevel near 1.42 volts). The waveform Comp_Out 230 toggles to a logiczero during transition 232, which reaches a logic zero level at time 202(e.g., around 10.33 microseconds).

At time 202, the simulation 200 approaches a steady-state response. Thewaveform INM 210 is around 2.92 volts and the waveform INP 220 is around2.90 volts. The waveform Source 240 is maintained at a voltage of around3.6 volts after the transition 242. The waveform Aux_bias 250 ismaintained at a voltage of around 3.90 volts because the feedbackcontrol of the replica input transistor pair 120 circuit is disabled.

The latency of the comparator 100 shown in the simulation 200 can bemeasured from the start of the transition 212 of the waveform INM 210 tothe end of the transition 232 of the waveform Comp_Out 230. When someasured, the latency of the comparator 100 in the simulation 200 isaround 330 nanoseconds, where the simulation 200 includes disabling thefeedback control of the replica input transistor pair 120 circuit. Inthe simulation described hereinbelow with respect to FIG. 3 (in whichthe feedback control of the replica input transistor pair 120 circuit isnot disabled as a simulation parameter), the latency of the comparator100 in the simulation 200 is reduced to around 124 nanoseconds.

FIG. 3 is a waveform diagram of an example simulation of a low-powerenhanced response of the example comparator to a large input signaltransition. The example simulation 300 includes waveforms for showing anexample operation of portions of the comparator 100, describedhereinabove with reference to FIG. 1. The example waveforms include: thewaveform INM 310, the waveform INP 320, the waveform Comp_Out 330, thewaveform Source 340, and the waveform Aux_bias 350. The low-powerenhanced response of the example comparator to a large input signaltransition is enabled in the simulation 300.

In the simulation 300 for illustrating a current-boost response of thecomparator 100 to a large input signal transition, the waveform INM 310is initially asserted at a ground potential (e.g., 0 volts) and thewaveform INP 320 is initially asserted at around 2.90 volts. Because themagnitude of the waveform INP 320 is greater than the magnitude of thewaveform INM 310 (e.g., under steady-state conditions), the waveformComp_Out 330 is initially a logic one (e.g., 1.42 volts).

The waveform Source 340 (Common_Source_Node) indicates the voltage ofthe common source node 114 is initially driven to around 1.2 volts inresponse to the current sourced by a first current source (e.g., currentsource I1), and in response the currents selectively controlled bytransistors Q1 and Q2. For example, the waveform Source 340 is driven toaround 1.2 volts in response to the waveform INM 310 being at groundpotential (e.g., which strongly biases Q2 to conduct), in response tothe waveform INP 320 being at around 2.90 volts (e.g., which moderatelybiases Q1 to conduct), and in response to the commonly controlledcurrent mirror of Q3 and Q4.

At 10 microseconds into the simulation 300, the waveform INM 310 isdriven to undergo a large voltage transition 312 from a ground potentialto a voltage around 2.92 volts. The transition 312 of waveform INM 310causes transient 322.

In response to the transition 312, the gate voltage of transistor Q1 israised to around 2.92 volts. Accordingly, the gate voltage of Q1 (aftertransition 312) is higher than the contemporaneous gate voltage of Q2.The low level of the current maximum of current source I1 can result ina current-starved response, which contributes to the latency of thevoltage rise (during transition 342) of the common source node 114. Thewaveform Source 340 during transition 342 is raised (e.g., slowly) inresponse to the limited current that is sourced by the current sourceI1, the common source node 114 capacitance, and the current drained bythe current mirror that includes transistors Q3 and Q4.

The replica input transistor pair 120 (e.g., being coupled to the inputsof the input transistor pair 112) detects the current-starved conditionof the common source node 114. In response to the current-starvedcondition, the tail current of the replica input transistor pair 120 isdecreased. In response to the decrease of the tail current of thereplica input transistor pair 120, less current is added to the currentflowing through the drain of Q9. In response to less current being addedto the current flowing through Q9, the voltage (waveform Aux_bias 350)of the source of Q8 falls (e.g., as transition 352). For example, thetransition 352 begins at around 40 nanoseconds after the beginning ofthe current-starved condition.

As the waveform Aux_bias 350 falls, the PMOS transistor (e.g., switch)Q7 increases conductivity and adds current (via the current switch 140)to the common source node 114. Adding current via Q7 of the currentswitch 140 to the common source node 114 reduces the current-starvedcondition at the common source node 114 and accelerates the rise of thewaveform Source 340 during transition 342.

As the waveform Source 340 rises to a steady-state level (e.g., 3.6volts) around the end of transition 342, the current-starved conditionof the common source node 114 is lessened. The replica input transistorpair 120 detects the reduction of the current-starved condition of thecommon source node 114, and the tail current of the replica inputtransistor pair 120 is increased. In response to the increase of thetail current of the replica input transistor pair 120, more current isadded to the current flowing through the drain of Q9. In response tomore current being added to the current flowing through Q9, the voltage(e.g., waveform Aux_bias 350) of the source of Q8 rises (e.g., astransition 354). For example, the transition 354 begins in response tothe waveform Source 340 rising to a steady-state level (e.g., whichoccurs at around 98 nanoseconds after the beginning of thecurrent-starved condition in the simulation 300).

As the waveform Aux_bias 350 rises during the transition 354, the PMOStransistor (e.g., switch) Q7 decreases conductivity and progressivelyadds less current (e.g., sourced from the current switch 140) to thecommon source node 114. The waveform Aux_bias 350 rises to asteady-state level (e.g., 3.6 volts) after the comparator 100 responds(e.g., correctly responds) to the relative change in the first inputsignal that occurs at 10 microseconds into the simulation 300.

In the simulation 300, the waveform Comp_Out 330 toggles in response tothe transition 312 of the waveform INM 310 to a voltage greater than thecontemporaneous voltage of the waveform INP 320. In response to thetransition 312, the waveform Comp_Out 330 toggles from a logic one to alogic zero. The waveform Comp_Out 330 toggles to a logic zero duringtransition 332, which reaches a logic zero level at time 302 (e.g.,around 10.124 microseconds).

After time 302, the simulation 300 approaches a steady-state response.The waveform INM 310 is around 2.92 volts and the waveform INP 320 isaround 2.90 volts. The waveform Source 340 reaches and is maintained ata steady-state voltage of around 3.60 volts after the transition 342.

The latency of the comparator 100 shown in the simulation 300 can bemeasured from the start of the transition 312 of the waveform INM 310 tothe end of the transition 332 of the waveform Comp_Out 330. When someasured, the latency of the comparator 100 in the simulation 300 isshortened to around 124 nanoseconds by the described current boost thatis added by the current switch 140. The latency of simulation 300 is 176nanoseconds faster than the latency of 330 nanoseconds of the simulation200, in which the feedback control of the replica input transistor pair120 circuit is disabled (e.g., in response to a simulation 200 parameterinput).

FIG. 4 is a waveform diagram of an example simulation of a disabledlow-power enhanced response of the example comparator to a small inputsignal transition. The example simulation 400 includes waveforms forshowing an example operation of portions of the comparator 100,described hereinabove with reference to FIG. 1. The example waveformsinclude the waveform INM 410, the waveform INP 420, the waveformComp_Out 430, the waveform Source 440, and the waveform Aux_bias 450.The low-power enhanced response of the example comparator to a smallinput signal transition can be disabled in response to a simulationparameter (e.g., a feedback signal to generate the low-power responsecan be disabled by coupling the Aux_bias node to 3.9 volts via an idealswitch in the simulation 400.

In the simulation 400 for illustrating a current-starved response of thecomparator 100 to a small input signal transition, the waveform INM 410is initially asserted (e.g., before time 402) at a potential of 2.88volts and the waveform INP 420 is initially asserted at around 2.90volts. Because the magnitude of the waveform INP 420 is greater than thecontemporaneous magnitude of the waveform INM 410 (e.g., understeady-state conditions), the waveform Comp_Out 430 is initially a logicone (e.g., 1.42V).

The waveform Source (Common_Source_Node) 440 indicates the voltage ofthe common source node 114 is initially driven to around 3.55 volts inresponse to the current sourced by a first current source (e.g., currentsource I1), and in response the currents selectively controlled bytransistors Q1 and Q2. For example, the waveform Source 440 is driven toaround 3.55 volts in response to the waveform INM 410 being at 2.88volts, in response to the waveform INP 420 being at around 2.90 volts,and in response to the commonly controlled current mirror of Q3 and Q4(e.g., which are commonly biased via the resistor network of R1 and R2).

The waveform Aux_bias 450 is driven to a value of around 3.90 volts inresponse to the current source I2. The low-power enhanced response ofthe example comparator to a small input signal transition can bedisabled in response to a simulation parameter (e.g., a feedback signalto generate the low-power response can be disabled by coupling theAux_bias node to 3.9 volts via an ideal switch in the simulation 400).

During operation of the comparator 100, the feedback signal of thereplica input transistor pair 120 circuit can be a voltage developed inresponse to the tail current of the replica input transistor pair 120circuit. In the simulation 400, the feedback signal is decoupled (e.g.,as a function of a simulation 400 input parameter) from modulating thewaveform Aux_bias 450. The feedback signal is disabled (e.g., by turningoff Q8 in response to a simulation 400 input parameter), so that, forexample, a response of the comparator 100 with a disabled low powerenhancement can be seen. Without the described low power selectedcurrent-boost enhancement being enabled in the example, the simulation400 of the comparator 100 shows a latency (e.g., around 40 nanoseconds)of the voltage (e.g., waveform Source 440) of the common source node 114rising to a steady-state level.

The waveform Aux_bias 450 is coupled to the control terminal (e.g.,gate) of the transistor Q7. The transistor Q7 is arranged as aprogrammable (e.g., programmable in response to a gate voltage) currentsource for selectively applying current to the common source node 114.In the simulation 400, the transistor Q7 is disabled (in response to asimulation 400 input) against selectively applying current to the commonsource node 114 (e.g., applying current in response to the feedbacksignal generated by the replica input transistor pair 120 circuit).Because the waveform Aux_bias 450 is around 3.90 volts, the PMOStransistor Q7 is in the off state, so that no boost current is injectedby the current source Q7 into the common source node 114 (of Q1 and Q2,for example).

At 10 microseconds into the simulation 400, the waveform INM 410 isdriven (e.g., as a simulation 400 input parameter) to undergo a smallvoltage transition 412 from a voltage of 2.88 volts to a voltage around2.92 volts. The transition 412 of waveform INM 410 causes transients 422and 452.

In response to the transition 412, the gate voltage of transistor Q2 israised to around 2.92 volts. Accordingly, the gate voltage of Q2 (aftertransition 412) is higher than the contemporaneous gate voltage of Q1.The current source I1 is designed to source current at a low maximum(e.g., to save power). The low level of the maximum current can resultin a current-starved response, which contributes to the latency of thevoltage rise (during transition 442) of the common source node 114. Thewaveform Source 440 during transition 442 is raised (e.g., slowly) inresponse to the limited current that is sourced by the current sourceI1, the common source node 114 capacitance, and the current drained bythe current mirror that includes transistors Q3 and Q4.

Parasitic conditions of the structures forming the common source node114 (e.g., of Q1 and Q2) impede a rise (e.g., instantaneous rise) in thevoltage of the common source node 114. Accordingly, the slew rate of thetransition 442 is limited, and the rise of the waveform Source 440 toaround 3.6 volts is achieved with a latency of around 40 nanoseconds.

In the simulation 400, the waveform Comp_Out 430 toggles in response tothe transition 412 of the waveform INM 410 to a voltage greater than thecontemporaneous voltage of the waveform INP 420. In response to thetransition 412, the waveform Comp_Out 430 toggles from a logic one to alogic zero. The waveform Comp_Out 430 toggles to a logic zero duringtransition 432, which reaches a logic zero level around 1 nanosecondbefore time 404 (e.g., where time 404 is 10.056 microseconds).

At time 404, the simulation 400 approaches a steady-state response. Thewaveform INM 410 is around 2.92 volts and the waveform INP 420 is around2.90 volts. The waveform Source 440 is maintained at a voltage of around3.57 volts after the transition 442. The waveform Aux_bias 450 ismaintained at a voltage of around 3.90 volts because the feedbackcontrol of the replica input transistor pair 120 circuit is disabled.

The latency of the comparator 100 shown in the simulation 400 can bemeasured from the start of the transition 412 of the waveform INM 410 tothe end of the transition 432 of the waveform Comp_Out 430. When someasured, the latency of the comparator 100 in the simulation 400 isaround 55 nanoseconds, where the simulation 400 includes disabling thefeedback control of the replica input transistor pair 120 circuit. Inthe simulation described hereinbelow with respect to FIG. 5 (in whichthe feedback control of the replica input transistor pair 120 circuit isnot disabled as a simulation parameter), the latency of the comparator100 in the simulation 500 is similar to the latency of the comparator100 in the simulation 400.

FIG. 5 is a waveform diagram of an example simulation of a low-powerenhanced response of the example comparator to a small input signaltransition. The example simulation 500 includes waveforms for showing anexample operation of portions of the comparator 100, describedhereinabove with reference to FIG. 1. The example waveforms include: thewaveform INM 510, the waveform INP 520, the waveform Comp_Out 530, thewaveform Source 540, and the waveform Aux_bias 550. The low-powerenhanced response of the example comparator to a small input signaltransition is enabled in the simulation 500. Under the initialconditions, the waveform Aux_bias is a logic one, so that the transistorQ7 is turned off, and so that Q7 does not (initially) inject a boostcurrent into the common source node 114.

In the simulation 500 for illustrating a current-boost response of thecomparator 100 to a small input signal transition, the waveform INM 510is initially asserted (e.g., before time 502) at a potential of 2.88volts and the waveform INP 520 is initially asserted at around 2.90volts. Because the magnitude of the waveform INP 520 is greater than themagnitude of the waveform INM 510 (e.g., under steady-state conditions),the waveform Comp_Out 530 is initially a logic one (e.g., 1.42 volts).

The waveform Source (Common_Source_Node) 540 indicates the voltage ofthe common source node 114 is initially driven (e.g., before time 502)to around 3.55 volts in response to the current sourced by a firstcurrent source (e.g., current source I1), and in response the currentsselectively controlled by transistors Q1 and Q2. For example, thewaveform Source 540 is driven to around 3.55 volts in response to thewaveform INM 510 being at 2.88 volts, in response to the waveform INP520 being at around 2.90 volts, and in response to the commonlycontrolled current mirror of Q3 and Q4.

At 10 microseconds into the simulation 500 (e.g., at time 502), thewaveform INM 510 is driven to undergo a small voltage transition 512from a voltage of 2.88 volts to a voltage around 2.92 volts. Thetransition 512 of waveform INM 510 causes transient 522.

In response to the transition 512, the gate voltage of transistor Q2 israised to around 2.92 volts. Accordingly, the gate voltage of Q2 (aftertransition 512) is higher than the contemporaneous gate voltage of Q1.The low level of the current maximum of current source I1 can result ina current-starved response, which contributes to the latency of thevoltage rise (during transition 542) of the common source node 114. Thewaveform Source 540 during transition 542 is raised in response to thelimited current that is sourced by the current source I1, the commonsource node 114 capacitance, and the current drained by the currentmirror that includes transistors Q3 and Q4.

The replica input transistor pair 120 (e.g., being coupled to the inputsof the input transistor pair 112) detects the current-starved conditionof the common source node 114. In response to the current-starvedcondition, the tail current of the replica input transistor pair 120 isdecreased. In response to the decrease of the tail current of thereplica input transistor pair 120, less current is added to the currentflowing through the drain of Q9. In response to less current being addedto the current flowing through Q9, the voltage (waveform Aux_bias 550)of the source of Q8 falls (e.g., as transition 552). For example, thetransition 552 begins at around 10 nanoseconds after the beginning ofthe current-starved condition.

As the waveform Aux_bias 550 falls, the PMOS transistor (e.g., switch)Q7 increases conductivity and adds current (from the current switch 140)to the common source node 114. In the simulation 500, the resultingcurrent-starved condition is small (e.g., because of the small voltagechange of the waveform INM 510, and the added current from the currentswitch Q7 to the common source node 114 does not readily noticeablyaccelerate (at the scale shown in FIG. 5) the rise of the waveformSource 540 during transition 542. As described hereinbelow, thetransition 552 does not readily noticeably turn on transistor Q7 foradding supplemental current to the common source node 114.

In the simulation 500, the waveform Comp_Out 530 toggles in response tothe transition 512 of the waveform INM 510 to a voltage greater than thecontemporaneous voltage of the waveform INP 520. In response to thetransition 512, the waveform Comp_Out 530 toggles from a logic one to alogic zero. The waveform Comp_Out 530 toggles to a logic zero duringtransition 532, which reaches a logic zero level at time 504 (e.g.,around 10.055 microseconds).

At time 504, the simulation 500 the waveform INM 510 is around 2.92volts, the waveform INP 520 is around 2.90 volts, and the waveformSource 540 reaches and is maintained at a steady-state voltage of around3.6 volts after the transition 542.

The latency of the comparator 100 shown in the simulation 500 can bemeasured from the start of the transition 512 of the waveform INM 510 tothe end of the transition 532 of the waveform Comp_Out 530. As shown inFIG. 5, the latency of the comparator 100 in the simulation 500 is canbe compared to the latency of the comparator 100 in the simulation 400(described hereinabove).

FIG. 6 is a waveform diagram of another example simulation of a disabledlow-power enhanced response of the example comparator to a large inputsignal transition. The example simulation 600 includes waveforms forshowing an example operation of portions of the comparator 100,described hereinabove with reference to FIG. 1. The example waveformsinclude the waveform INM 610, the waveform INP 620, the waveformComp_Out 630, the waveform Source 640, the waveform Aux_bias 650, thewaveform 1st_stage_out_plus 660, the waveform 1st_stage_out_minus 670,and the waveform 2nd_stage_output 680. The transistor Q8 is turned offby a simulation 600 parameter to disable a feedback signal (e.g., viathe waveform Aux_bias 650) for controlling the current boosting of Q7.

The simulation 600 shows a current-starved response of a common sourcenode 114 of the comparator 100 to a first input signal change to a firstvalue that is substantially close to a second value of a second inputsignal. For example, a first input signal value is substantially closeto a second input signal value when the difference therebetween is avalue within the input offset of the amplifier that receives the firstand second input signals.

In the simulation 600, the waveform INM 610 is initially asserted (e.g.,before the 10-microsecond mark of the simulation 600) at a groundpotential (e.g., 0 volts), whereas the waveform INP 620 is initiallyasserted at around 2.90 volts. Because the initial magnitude of thewaveform INP 620 is greater than the contemporaneous magnitude of thewaveform INM 610 (e.g., under initial steady-state conditions), thewaveform Comp_Out 630 is initially a logic one (e.g., 1.42V). Thewaveform Comp_Out 630 is generated in response to the difference betweenthe waveform 1st_stage_out_minus 670 and the waveform 1st_stage_out_plus660, where the difference is indicated by the waveform 2nd_stage_output680 described hereinbelow.

The waveform Source (Common_Source_Node) 640 indicates the voltage ofthe common source node 114 is initially driven to a low voltage (e.g.,1.2 volts) in response to the current sourced by a first current source(e.g., current source I1), and in response the currents selectivelycontrolled by transistors Q1 and Q2. For example, the waveform Source640 is driven to a low voltage (e.g., 1.2 volts) in response to thewaveform INM 610 being at ground, in response to the waveform INP 620being at around 2.90 volts, and in response to the commonly controlledcurrent mirror of Q3 and Q4 (e.g., which are commonly biased via theresistor network of R1 and R2).

The waveform Aux_bias 650 is driven to a value of around 3.90 volts inresponse to the current source I2. The low-power enhanced response ofthe example comparator to a large input signal transition can bedisabled in response to a simulation parameter (e.g., a feedback signalto generate the low-power response can be disabled by coupling theAux_bias node to 3.9 volts via an ideal switch in the simulation 600).

During operation of the comparator 100, the feedback signal of thereplica input transistor pair 120 circuit can be a voltage developed inresponse to the tail current of the replica input transistor pair 120circuit. In the simulation 600, the feedback signal is decoupled (e.g.,in response to a simulation 600 input parameter) from the waveformAux_bias 650. The feedback signal is disabled (e.g., in response to asimulation 600 input parameter), so that, for example, a response of thecomparator 100 with a disabled low power enhancement can be seen. Forexample, a deficit of “robust” operation (e.g., a susceptibility togenerating erroneous outputs) can be demonstrated by the examplespurious output pulse of the waveform Comp_Out 630 (e.g., where thespurious pulse includes transitions 632 and 634).

The waveform Aux_bias 650 is coupled to the control terminal (e.g.,gate) of the transistor Q7. The transistor Q7 is arranged as aprogrammable (e.g., programmable in response to a gate voltage) currentsource for selectively applying current to the common source node 114.In the simulation 600, the transistor Q7 is disabled (in response to asimulation 600 input) against selectively applying current to the commonsource node 114 (e.g., applying current in response to the feedbacksignal generated by the replica input transistor pair 120 circuit).Because the waveform Aux_bias 650 is around 3.90 volts throughout thesimulation 600, the PMOS transistor Q7 is in the off state, so that noboost current is injected by the current source Q7 into the commonsource node 114 (of Q1 and Q2, for example).

The waveform 1st_stage_out_plus 660 and the waveform 1st_stage_out_minus670 are developed in response to the first input signal and the secondinput signal, respectively. For example, the waveform 1st_stage_out_plus660 is initially around 169 millivolts (in response to the waveform INM610), and the waveform 1st_stage_out_minus 670 is around 550 millivolts(in response to the waveform INP 620).

The second stage 150 generates the waveform 2nd_stage_output 680 inresponse to the difference between the waveform 1st_stage_out_plus 660and the waveform 1st_stage_out_minus 670 (where the waveform1st_stage_out_plus 660 and the waveform 1st_stage_out_minus 670 arefirst stage 110 output signals). The difference between the initialvoltage values of the waveform 1st_stage_out_plus 660 and the waveform1st_stage_out_minus 670 is greater than an input offset of the secondstage 150 input transistors Q12 and Q13 (e.g., which entails a reducedprobability of an output error). When the difference between the initialvoltage values of the waveform 1st_stage_out_plus 660 and the waveform1st_stage_out_minus 670 is not greater than an input offset of thesecond stage 150, the second stage is susceptible to outputting anerroneous output (as described hereinbelow).

In response to the difference of the input signals, the waveform2nd_stage_output 680 is initially a voltage (e.g., a ground or anear-ground potential) for indicating that the magnitude of the waveform1st_stage_out_plus 660 is less than the waveform 1st_stage_out_minus670. The waveform 2nd_stage_output 680 value (e.g., initially at groundpotential) is received by the third stage 160 and is quantized by aninput gate of the third stage 160 (e.g., initially as an input logiczero). The third stage 160 buffers and inverts the input logic zero andoutputs the buffered inverted value as the waveform Comp_Out 630 (e.g.,which initially is a logic one).

At 10 microseconds into the simulation 600, the waveform INM 610 isdriven (e.g., as a simulation 600 input parameter) to undergo a largevoltage transition 612 from a ground potential to a voltage around 2.88volts. The rise of the waveform INM 610 to the voltage around 2.88 voltsis a level that continues to be less than the magnitude of the waveformINP 620 (e.g., 2.90 volts), so the waveform Comp_Out 630 is ideally(e.g., without logical error) expected to not toggle (e.g., changeoutput logic state). The transition 612 of waveform INM 610 causestransients 622, 652, 662, 672, and 682.

During the transition 612, the gate voltage of transistor Q2 is raisedto around 2.88 volts. After the transition 612, the gate voltage of Q2remains lower than the contemporaneous gate voltage of Q1 (e.g., so thatthe waveform Comp_Out 630 does not properly toggle). The current sourceI1 is designed to source current to the Common_Source_Node (e.g.,waveform Source 640) at a low maximum current (e.g., by design, to savepower). The low level of the maximum current results in acurrent-starved response (e.g., in response to the transition 612),which contributes to the latency of the voltage rise of transition 642.

The current-starved-induced latency (e.g., which occurs duringtransition 642) of the common source node 114 of the first stage 110also contributes to the latency of the settling of the waveforms1st_stage_out_plus 660 and the 1st_stage_out_minus 670. For example,after the transients 662 and 672, the voltages of the waveforms1st_stage_out_plus 660 and the 1st_stage_out_minus 670 converge to adifference of less than an input offset of the second stage 150amplifier (which can lead to an erroneous output of the second stage150).

The convergence of the waveform 1st_stage_out_plus 660 and the1st_stage_out_minus 670 to low voltages (e.g., near ground) helps ensurethat the NMOS transistors Q12 and Q13 are more negatively biased, sothat the transistors Q12 and Q13 do not conduct strongly. When thetransistors Q12 and Q13 do not conduct strongly during thecurrent-starved condition (e.g., during transition 642), the waveform2nd_stage_output 680 rises (e.g., gradually rises in response to a draincurrent generated by the current mirror formed by the transistors Q10and Q11).

As the waveform 2nd_stage_output 680 rises in the simulation 600, thesecond stage 150 output (logically erroneously) reaches a voltage (e.g.,900 millivolts) that can be quantized by the third stage 160 as a logicone. The third stage 160 inverts the received logic one value, whichcauses the third stage 160 to toggle low at transition 632. Thetransition of the third stage 160 output to toggle is erroneous becausethe input signal having the greatest magnitude of the first and secondinput signals (e.g., the waveform INM 610 and the waveform INP 620) hasnot changed. Such errors can lead to incorrect processing of data, sothat the incorrect processing can result in corrupted output data orother processing errors. When the comparator is used in a feedback loopfor controlling an external (e.g., to the comparator 100) process (forexample), the external process can interrupt the stability and accuracyof the feedback control signals (including signals withinsafety-critical systems).

The waveform Source 640 reaches a steady-state value (e.g., nearsteady-state value) at the end of transition 642. At the end oftransition 642 (e.g., at time 602), the current-starved condition isalleviated, so that the waveform 1st_stage_out_plus 660 and the waveform1st_stage_out_minus 670 rise. As the waveform 1st_stage_out_plus 660 andthe waveform 1st_stage_out_minus 670 rise, at least one of thetransistors Q12 and Q13 is biased more strongly (e.g., to increaseconductivity). As at least one of the transistors Q12 and Q13 is biasedmore strongly, the output (e.g., waveform 2nd_stage_output 680) of thesecond stage 150 falls to a level that can be quantized by the thirdstage 160 as a logic zero. The third stage 160 inverts the receivedlogic zero value, which causes the third stage 160 to toggle high (e.g.,back to a logically correct value) at transition 634. The transition 634of the third stage 160 output restores the correct output value of thecomparator 100 (e.g., after the current-starved condition duringtransition 642 is alleviated).

At time 604, the simulation 600 approaches a steady-state response. Thewaveform INM 610 is around 2.88 volts and the waveform INP 620 is around2.90 volts. The waveform Source 640 is maintained at a voltage of around3.44 volts after the transition 642. The waveform Aux_bias 650 ismaintained at a voltage of around 3.90 volts because the feedbackcontrol of the replica input transistor pair 120 circuit is disabled.The waveform 1st_stage_out_plus 660 is around 332 millivolts, thewaveform 1st_stage_out_minus 670 is around 394 millivolts, and thewaveform 2nd_stage_output 680 is around 217 millivolts.

The latency of the comparator 100 during transition 642 can result inerroneous performance (e.g., a logically incorrect output value) duringthe current-starved condition encountered, for example when at least oneinput signal undergoes a large transition (while not transitioning to avalue for indicating a logically valid transition of the comparator100). The simulation 600 includes disabling the feedback control of thereplica input transistor pair 120 circuit, so that the logicallyerroneous behavior of a comparator under a current-starved condition canbe demonstrated. In the simulation 700 (in which the feedback control ofthe replica input transistor pair 120 circuit is not disabled as asimulation parameter), the comparator 100 selectively couples currentfrom a current switch (e.g., current switch 140) into the common sourcenode 114 (e.g., to more quickly alleviate the current-starved conditionwhich can otherwise result in the generation of an erroneous outputsignal).

FIG. 7 waveform diagram of another example simulation of an enabledlow-power enhanced response of the example comparator to a large inputsignal transition. The example simulation 700 includes waveforms forshowing an example operation of portions of the comparator 100,described hereinabove with reference to FIG. 1. The example waveformsinclude the waveform INM 710, the waveform INP 720, the waveformComp_Out 730, the waveform Source 740, the waveform Aux_bias 750, thewaveform 1st_stage_out_plus 760, the waveform 1st_stage_out_minus 770,and the waveform 2nd_stage_output 780. The low-power enhanced responseof the example comparator to a large input signal transition is enabledin the simulation 700.

The simulation 700 shows a current-starved response of the comparator100 to a voltage transition of a first input signal to a first inputsignal value within a difference given by the input offset of theamplifier that receives the first and second input signals.

In the simulation 700, the waveform INM 710 is initially asserted (e.g.,before the 10-microsecond mark of the simulation 700) at a groundpotential, whereas the waveform INP 720 is initially asserted at around2.90 volts. Because the initial magnitude of the waveform INP 720 isgreater than the contemporaneous magnitude of the waveform INM 710(e.g., under initial steady-state conditions), the waveform Comp_Out 730is initially a logic one (e.g., 1.42V). The waveform Comp_Out 730 isgenerated in response to the difference between the waveform1st_stage_out_minus 770, and the waveform 2nd_stage_output 780 describedhereinbelow.

The waveform Source (Common_Source_Node) 740 indicates the voltage ofthe common source node 114 is initially driven to a low voltage (e.g.,1.2 volts) in response to the current sourced by a first current source(e.g., current source I1), and in response the currents selectivelycontrolled by transistors Q1 and Q2. For example, the waveform Source740 is driven to a low voltage (e.g., 1.2 volts) in response to thewaveform INM 710 being at ground, in response to the waveform INP 720being at around 2.90 volts, and in response to the commonly controlledcurrent mirror of Q3 and Q4.

The waveform Aux_bias 750 is driven to a value of around 3.90 volts inresponse to the current source I2. The waveform Aux_bias is not affectedby the Replica_load feedback signal of the replica input transistor pair120 circuit because the replica input transistor pair 120 circuit doesnot (under initial conditions) detect a current starved condition of thecommon source node 114. The transistor Q8 is biased to be on in responseto the signal NCAS (“normalize cascode,” shown in FIG. 1). When thetransistor Q8 is in the on state, the feedback path (e.g., includingsignal Replica_load, the first and second current terminals oftransistor Q8, and the signal Aux_bias) of the feedback signal of thereplica input transistor pair 120 circuit are coupled as a feedbackcircuit (e.g., for coupling a feedback signal to the common source node114).

During operation of the comparator 100, the feedback signal of thereplica input transistor pair 120 circuit can be a voltage developed inresponse to the tail current of the replica input transistor pair 120circuit. In the simulation 700, the feedback signal is coupled to thegate of transistor Q7 (and to the common source node 114 via the drainof Q7), so that, for example, a response of the replica input transistorpair 120 can be seen. For example, a “robust” operation (e.g., aninsusceptibility to generating erroneous outputs from input signalshaving a voltage input less than an input offset of an amplifier) can bedemonstrated by the lack of a spurious output pulse (cf., Comp_Out 630)of the waveform Comp_Out 730.

The waveform Aux_bias 750 is coupled to the control terminal (e.g.,gate) of the transistor Q7. In the simulation 700, the transistor Q7 isenabled (in response to a simulation 700 input) to selectively applycurrent to the common source node 114 (e.g., in response to the feedbacksignal generated by the replica input transistor pair 120 circuit).Because the waveform Aux_bias 750 is initially around 3.90 volts in thesimulation 700, the PMOS transistor Q7 is in the off state, so that noboost current is injected by the current source Q7 into the commonsource node 114 (of Q1 and Q2, for example).

The waveform 1st_stage_out_plus 760 and the waveform 1st_stage_out_minus770 are developed in response to the first input signal and the secondinput signal, respectively. For example, the waveform 1st_stage_out_plus760 is initially around 169 millivolts (in response to the waveform INM710), and the waveform 1st_stage_out_minus 770 is around 550 millivolts(in response to the waveform INP 720).

In response to the difference of the input signals, the waveform2nd_stage_output 780 initially indicates the magnitude of the waveform1st_stage_out_plus 760 is less than the waveform 1st_stage_out_minus770. The waveform 2nd_stage_output 780 value (e.g., initially at groundpotential) is received by the third stage 160 and is quantized by aninput gate of the third stage 160 (e.g., initially as an input logiczero). The third stage 160 buffers and inverts the input logic zero andoutputs the buffered inverted value as the waveform Comp_Out 730 (e.g.,which initially is a logic one).

At 10 microseconds into the simulation 700, the waveform INM 710 isdriven (e.g., as a simulation 700 input parameter) to undergo a largevoltage transition 712 from a ground potential to a voltage around 2.88volts. The rise of the waveform INM 710 to the voltage around 2.88 voltsis a level that continues to be less than the magnitude of the waveformINP 720, so the waveform Comp_Out 730 ideally (e.g., without error) doesnot toggle. The transition 712 of waveform INM 710 causes transients722, 762, 772, and 782.

During the transition 712, the gate voltage of transistor Q1 is raisedto around 2.88 volts. After the transition 712, the gate voltage of Q1remains lower than the contemporaneous gate voltage of Q2 (e.g., so thatthe waveform Comp_Out 730 does not properly toggle). The current sourceI1 is designed to source current at a low maximum (e.g., by design, tosave power), so that a current-starved response of the first stage 110is developed.

The current-starved-induced latency (e.g., which occurs duringtransition 742) of the common source node 114 of the first stage 110also contributes to the latency of the settling of the waveforms1st_stage_out_plus 760 and the 1st_stage_out_minus 770. For example,after transients 762 and 772, the voltages of the waveforms1st_stage_out_plus 760 and the 1st_stage_out_minus 770 converge to adifference of less than an input offset of the second stage 150amplifier (which can otherwise lead to an erroneous output of the secondstage 150).

The convergence of the waveform 1st_stage_out_plus 760 and the1st_stage_out_minus 770 to low voltages (e.g., near ground) helps ensurethat the NMOS transistors Q12 and Q13 are more negatively biased, sothat the transistors Q12 and Q13 do not conduct strongly. When thetransistors Q12 and Q13 do not conduct strongly during thecurrent-starved condition (e.g., during transition 742), the waveform2nd_stage_output 780 rises (e.g., gradually rises in response to a draincurrent generated by the current mirror formed by the transistors Q10and Q11).

As the second stage 150 output (e.g., the waveform 2nd_stage_output 780)rises in the simulation 700, the second stage 150 output is preventedfrom reaching a voltage that can be quantized by the third stage 160 asa logic one. As described herein following, the second stage 150 outputis prevented from reaching a voltage that would otherwise be quantizedby the third stage 160 as a logic one. The rise to the logic onethreshold of the second stage 150 output is prevented by the currentinjection selectively coupled via the current switch 140 (where thecurrent injection alleviates the current-starved condition of the commonsource node 114).

The replica input transistor pair 120 (e.g., being coupled to the inputsof the input transistor pair 112) detects the current-starved conditionof the common source node 114. In response to the current-starvedcondition, the tail current of the replica input transistor pair 120 isdecreased. In response to the decrease of the tail current of thereplica input transistor pair 120, less current is added to the currentflowing through the drain of Q9. In response to less current being addedto the current flowing through Q9, the voltage (waveform Aux_bias 750)of the source of Q8 falls (e.g., as transition 752). For example, thetransition 752 begins at around 40 nanoseconds after the beginning ofthe current-starved condition.

As the waveform Aux_bias 750 falls, the PMOS transistor (e.g., switch)Q7 increases conductivity and adds current (selectively sourced via thecurrent switch 140) to the common source node 114. Adding current viathe current switch Q7 to the common source node 114 reduces thecurrent-starved condition at the common source node 114 and acceleratesthe rise of the waveform Source 740 during transition 742.

As the waveform Source 740 rises to a steady-state level (e.g., 3.6volts) around the end of transition 742, the current-starved conditionof the common source node 114 is lessened. The replica input transistorpair 120 detects the reduction of the current-starved condition of thecommon source node 114, and the tail current of the replica inputtransistor pair 120 is increased. In response to the increase of thetail current of the replica input transistor pair 120, more current isadded to the current flowing through the drain of Q9. In response tomore current being added to the current flowing through Q9, the voltage(e.g., waveform Aux_bias 750) of the source of Q8 rises (e.g., astransition 754). For example, the transition 754 begins in response tothe waveform Source 740 rising to a steady-state level (e.g., whichoccurs at around 98 nanoseconds after the beginning of thecurrent-starved condition in the simulation 700).

As the waveform Aux_bias 750 rises during the transition 754, the PMOStransistor (e.g., switch) Q7 decreases conductivity and progressivelyadds less current (from the current switch 140) to the common sourcenode 114. The waveform Aux_bias 750 rises to a steady-state level (e.g.,3.6 volts) after the comparator 100 responds (e.g., correctly responds)to the relative change in the first input signal that occurs at 10microseconds into the simulation 700.

At time 704, the simulation 700 approaches a steady-state response. Thewaveform INM 710 is around 2.88 volts and the waveform INP 720 is around2.90 volts. The waveform Source 740 is maintained at a voltage of around3.44 volts after the transition 742. The waveform Aux_bias 750 ismaintained at a voltage of around 3.90 volts because the current switchQ7 is off. The waveform 1st_stage_out_plus 760 is around 331 millivolts,the waveform 1st_stage_out_minus 770 is around 392 millivolts, and thewaveform 2nd_stage_output 780 is around 217 millivolts.

In the simulation 700 of the comparator 100, the feedback control of thereplica input transistor pair 120 circuit detects the current-starvedcondition of the common source node 114. In response to an indication ofthe current-starved condition generated by the replica input transistorpair 120 circuit, the current switch 140 selectively couples currentfrom the current switch 140 into the common source node 114, whichaccelerates the reduction of the current-starved condition of the commonsource node 114. Accelerating the reduction of the current-starvedcondition of the common source node 114 prevents the convergence of thewaveforms 1st_stage_out_plus 760 and the 1st_stage_out_minus 770 to adifference of less than an input offset of the second stage 150amplifier. Inhibiting the convergence of the waveforms1st_stage_out_plus 760 and the 1st_stage_out_minus 770 converge to adifference of less than an input offset of the second stage 150amplifier increases the robustness of the comparator against generatingerroneous output signals.

FIG. 8 is a schematic diagram of another example Aux_bias generator forlow-power enhanced response of the example comparator. The exampleAux_bias generator 830 of circuit 800 includes transistors Q80, Q81,Q82, and Q83 and inverter 832. For example, the Aux_bias generator 830is similar to the Aux_bias generator 130 described hereinabove.

In at least one implementation, the PMOS transistors Q80 and Q81 arearranged as a current mirror, in which the current flowing through Q80controls the current flowing through Q81. The current flowing throughQ80 is controlled by the bias signal nbias, so that the NMOS transistorQ81 is biased to conduct (e.g., opens a channel for carrying) a firstcurrent “A” of magnitude “x” (e.g., xA). The Replica_load_CG signal(e.g., the gate control signal of Q3 and Q4 generated by the first stage110) is coupled to bias the NMOS transistor Q83 in response to thereplica input transistor pair 120 circuit for detecting acurrent-starved condition.

When the Replica_load_CG signal indicates that a current-starvedcondition does not exist, the transistor Q83 is biased to conduct acurrent (1.5×A) that is 50 percent larger than, for example, the currentsupplied by the transistor Q81. Because the biased current capacity ofQ83 is larger than the biased current capacity of Q81, the voltagedeveloped between the respective drains of Q81 and Q83 is quantized as alogical zero by the inverter 832. In response, the inverter 832 outputsa logical one, so that the PMOS transistor Q7 is turned off and thecurrent switch 140 does not inject additional current into the commonsource node 114.

When the Replica_load_CG signal indicates that a current-starvedcondition exists, the transistor Q83 is biased to not conduct (and/or toconduct a current that is of a small magnitude that causes the inverter832 to toggle, for example). When the transistor Q83 is biased to notconduct, the voltage developed between the respective drains of Q81 andQ83 is quantized as a logical one by the inverter 832. In response, theinverter 832 outputs a logical zero, so that the PMOS transistor Q7 isturned on and the current switch 140 is coupled to inject additionalcurrent into the common source node 114 (e.g., so the current-starvedcondition of the common source node 114 is alleviated).

For example, the Replica_loadz_CG signal can indicate that acurrent-starved condition exists by a reducing the voltage level of theReplica_load_CG signal, (e.g., so that the NMOS transistor Q83 isarranged to conduct less and/or to be turned off). When the voltage ofthe Replica_load signal is reduced to a voltage level selected toindicate that a current-starved condition exists, the Aux_bias generator830 is coupled to assert the bias signal to activate the current switch140 (e.g., in response to a decrease in voltage of the indicationreceived from the Replica_load_CG signal node). Accordingly, the currentof the replica load signal can be reduced in response to the voltagedrop of the common source node.

FIG. 9 is a flow diagram of an example method for a response of anexample low power comparator to input signal fluctuations. The process910 of the example method 900 comprises generating, by a first stage, afirst stage output signal in response to an input signal, wherein theinput signal is coupled to control a first current coupled from a firstcurrent source through a common node to generate the first stage outputsignal.

The process 920 comprises generating, by a replica input transistorpair, a replica load signal in response to the input signal and inresponse to current received from the common node. In an example, thereplica load signal is generated in response to detectingcurrent-starved response of an input transistor pair arranged togenerate the first stage output signal. In another example, the replicaload signal is generated in response to an emulation of the inputtransistor pair.

The process 930 comprises selectively coupling a second current from asecond current source to the common node in response to the replica loadsignal.

Various examples (and example operations thereof) of a comparator havinglower power consumption and reduced latencies are described herein withrespect to the accompanying FIGURES. A figure of merit of the lowerpower consumption and reduced latencies can be the multiplicativeproduct of the power consumed and the reduction in latency (e.g.,power*delay). In various simulations described hereinabove, athree-times improvement in comparator delay for same power have beenobserved. The synergy of the increase in robustness with the reducedpower and latencies (e.g., described hereinabove with respect tosimulation 700) can extend the figure of merit beyond three times insituations in which the input signals are within narrow margins.

Modifications are possible in the described examples, and other examplesare possible, within the scope of the claims.

What is claimed is:
 1. A circuit comprising: a differential input thatincludes a first input and a second input; an amplifier stage thatincludes a first set of input transistors coupled to a common node,wherein the first set of input transistors includes a first transistorthat includes a gate coupled to the first input, and a second transistorthat includes a gate coupled to the second input; a second set of inputtransistors coupled between the common node and a replica node, whereinthe second set of input transistors includes a third transistor thatincludes a gate coupled to the first input, and a fourth transistor thatincludes a gate coupled to the second input; a bias generator coupled tothe replica node and a bias node; and a current switch that includes afifth transistor that includes a gate coupled to the bias node, whereinthe fifth transistor is coupled to the common node.
 2. The circuit ofclaim 1, wherein the bias generator includes: a current source coupledto the bias node; a sixth transistor coupled between the bias node andthe replica node; and a seventh transistor coupled between the replicanode and a ground node.
 3. The circuit of claim 1, wherein the currentswitch includes a resistor coupled between a voltage node and the fifthtransistor.
 4. The circuit of claim 1, wherein: the amplifier stagefurther includes a differential output that includes a first output anda second output; the first transistor is coupled between the common nodeand the first output; and the second transistor is coupled between thecommon node and the second output.
 5. The circuit of claim 4, whereinthe amplifier stage further includes: a first resistor coupled betweenthe first output and a divided voltage node; and a second resistorcoupled between the second output and the divided voltage node.
 6. Thecircuit of claim 5, wherein the amplifier stage further includes: asixth transistor coupled between the first output and a ground node; anda seventh transistor coupled between the second output and the groundnode.
 7. The circuit of claim 6, wherein: the six transistor includes agate coupled to the divided voltage node; and the seventh transistorincludes a gate coupled to the divided voltage node.
 8. The circuit ofclaim 4, wherein: the amplifier stage is a first amplifier stage; andthe circuit further comprises a second amplifier stage coupled to thedifferential output of the first amplifier stage.
 9. The circuit ofclaim 8, wherein the second amplifier stage further includes: a sixthtransistor that includes a gate coupled to the first output of the firstamplifier stage; and a seventh transistor that includes a gate coupledto the second output of the first amplifier stage.
 10. The circuit ofclaim 9, wherein the second amplifier stage further includes: an eighthtransistor coupled between a voltage node and the sixth transistor; anda ninth transistor coupled between the voltage node and the seventhtransistor.
 11. The circuit of claim 1, further comprising a currentsource coupled to the common node.
 12. A circuit comprising: anamplifier stage that includes a first set of input transistors coupledto a common node; a replica input transistor pair coupled to the commonnode and configured to: detect a voltage drop on the common node; andprovide a replica signal based on the voltage drop on the common node; abias generator coupled to the replica input transistor pair andconfigured to: receive the replica signal; and provide a bias signalbased on the replica signal; and a current switch coupled to the biasgenerator and to the common node and configured to: receive the biassignal, and provide current to the common node based on the bias signal.13. The circuit of claim 12 further comprising a differential input,wherein: the differential input includes a first input and second input;the first set of input transistors includes: a first transistor thatincludes a gate coupled to the first input; and a second transistor thatincludes a gate coupled to the second input; and the replica inputtransistor pair includes: a third transistor that includes a gatecoupled to the first input; and a fourth transistor that includes a gatecoupled to the second input.
 14. The circuit of claim 13, wherein eachof the first, second, third, and fourth transistors is coupled to thecommon node.
 15. The circuit of claim 13, wherein the third and fourthtransistors are coupled between the common node and a replica node toprovide the replica signal at the replica node.
 16. The circuit of claim15, wherein the bias generator includes: a current source; a fifthtransistor coupled between the current source and the replica node; anda sixth transistor coupled between the fifth transistor and a groundnode.
 17. The circuit of claim 16, wherein the current switch includes:a resistor coupled to a voltage node; and a seventh transistor coupledbetween the resistor and the common node, wherein the seventh transistorincludes a gate coupled to the current source and to the fifthtransistor.
 18. The circuit of claim 13, wherein the amplifier stagefurther includes: a first resistor coupled between the first transistorand a divided voltage node; and a second resistor coupled between thesecond transistor and the divided voltage node.
 19. The circuit of claim18, wherein the amplifier stage further includes: a fifth transistorcoupled between the first transistor and a ground node; and a sixthtransistor coupled between the second transistor and the ground node.20. The circuit of claim 12, further comprising a current source coupledto the common node.